Signal reception and processing method for cordless communications systems

ABSTRACT

Channel selection of a received signal is first of all carried out, in the process, by way of an analog channel selection filter. The signal is then converted to a digital discrete-time and discrete-value signal. Finally, the continuous-time and continuous-value signal profile is determined on the basis of a mathematical reconstruction using the zero crossings {t i } and the phase values {φ(t i )=k i   ·π/2 , k i εN 0 }, with a mathematical reconstruction algorithm using a function system {φ(t−k)}.

CROSS-REFERENCE TO RELATED APPLICATION

[0001] This application is a continuation of copending InternationalApplication No. PCT/DE02/00017, filed Jan. 7, 2002, which designated theUnited States and which was not published in English.

BACKGROUND OF THE INVENTION Field of the Invention

[0002] The present invention relates to a method for processing areceived signal in a cordless communications system, in particular for acordless telephone, and to a receiver circuit which operates using themethod.

[0003] Cordless digital communications systems such as DECT, WDCT,Bluetooth, SWAP, WLAN IEEE802.11 require suitable receivers, whichsupply the demodulator with a baseband signal with as little distortionas possible in a simple manner, for wire-free reception of theradio-frequency signals that are transmitted via the air interface. Inaddition to high sensitivity, a high degree of integration, low costs,low power consumption as well as flexibility in terms of theapplicability to different digital communications systems are desirablein this case. In order to exploit the advantages of digital circuittechnology (no drift, no aging, no temperature dependency, exactreproducibility), at least a portion of the receiver circuit is in thiscase in the form of digital signal processing elements. In this case,signal distortion can occur not only in the analog signal processingsection (so-called analog receiver front end) but also in the digitalsignal processing section, and the characteristics of this signaldistortion depend on the (analog and digital) signal processing elementsthat are used. Signal distortion such as this reduces the powerefficiency of the receiver, that is to say it adversely affects thesensitivity and the range of the receiver for a predetermined bit errorrate.

[0004] Superheterodyne receivers are currently frequently used forcordless digital communications systems. In order to achieve greatersystem integration and thus lower system costs, the receivers with a lowintermediate frequency are thus also increasingly being used, so-calledlow-IF (intermediate frequency) receivers or zero-IF (homodyne)receivers, since they do not require any external filters for mirrorfrequency suppression and thus allow greater system integration (see,for example, DECT, Bluetooth, WDCT). Currently, analog FM demodulators(frequency modulation) based on the limiter/discriminator principle areused on the basis of the digital modulation GFSK which is used in thecordless systems and for which a formulation on the basis of frequencymodulation is possible. The limiter is followed by analogfrequency-selective filtering in order to suppress the relatively highfrequency interference that is caused by the nonlinearity of thelimiter. From a signal theory point of view, this filtering is notoptimum since, even if the signal that is modulated onto theintermediate frequency is band-limited exactly and the instantaneousphase φ(t) is band-limited exactly, the complex envelope e^(iφ(t)) whichis subjected to the filtering process is not band-limited exactly.

SUMMARY OF THE INVENTION

[0005] It is accordingly an object of the invention to provide a methodfor processing a received signal in a cordless communications system,and a corresponding receiver circuit which overcome the above-mentioneddisadvantages of the heretofore-known devices and methods of thisgeneral type and which allow improved signal processing from thesignal-theory point of view, in particular for signals that aremodulated using digital signal transmission methods such as FSK(Frequency Shift Keying).

[0006] With the foregoing and other objects in view there is provided,in accordance with the invention, a method for processing a signal, suchas a digitally modulated signal in a cordless communications system. Themethod comprises the following steps:

[0007] carrying out channel selection on a received signal with ananalog channel selection filter;

[0008] converting the signal to a digital, discrete-time anddiscrete-value signal;

[0009] mathematically reconstructing a continuous-time andcontinuous-value signal profile using zero crossings {t_(i)} and phasevalues {φ(t_(i))=k_(i)·π/2, k_(i)εN₀} by way of a mathematicalreconstruction algorithm using a function system {φ(t−k)}.

[0010] The invention is primarily based on the idea that, after carryingout channel selection for the received signal, the signal is convertedto a digital, discrete-time and discrete-value signal, and amathematical reconstruction of the signal profile is then carried out onthe basis of the zero crossings of the complex envelope, by means of amathematical reconstruction algorithm using a function system.

[0011] In other words, the method according to the invention forprocessing a received signal in a cordless communications system has thefollowing steps:

[0012] channel selection is carried out by means of an analog channelselection filter (KSF);

[0013] the signal is converted to a digital, discrete-time anddiscrete-value signal;

[0014] the continuous-time and continuous-value signal profile ismathematically reconstructed using the zero crossings {t_(i)} and thephase values {φ(t_(i))=k_(i)·π/2, k_(i)εN₀} by means of a mathematicalreconstruction algorithm using a function system {φ(t−k)}.

[0015] In one embodiment of a digital receiver, a frequency conversionis carried out to an intermediate frequency. Thus, in comparison to theknown solutions, the method proposed here makes use of the fact that adiscrete-value (binary) complex signal is produced after the limiter,whose useful information is contained in the zero crossings of the I andQ, or real and imaginary part. Since this signal is initially stillcontinuous in time, the change to a digital (discrete-time anddiscrete-value) signal is carried out by means of equidistant samplingat a sampling rate f_(s). The mathematical reconstruction of theinstantaneous phase φ(t) of the signal is carried out purely digitally,exclusively using the zero crossings and the phase values, which can bedetermined if the intermediate frequency is chosen suitably,corresponding to the reconstruction algorithm which is described in moredetail below.

[0016] By way of example, shifted orthogonal sinc functions (see theShannon-Whittaker sampling theorem) or orthogonal scaling functions (seewavelets) can be used for the function system {φ(t−k)}, depending on thecharacteristics of the signal s(t) to be reconstructed. The so-calledDaubechies scaling functions may be mentioned as an example for thispurpose.

[0017] However, the method according to the invention may also be usedfor nonorthogonal function systems, for example bi-orthogonal functionsystems.

[0018] In order to improve the already achieved signal quality and noisefiltering even further, it is possible to carry out filtering subsequentto the mathematical reconstruction, so-called postfiltering by means ofa digital filter with a predetermined system function.

[0019] The method according to the invention is particularly suitablefor reconstruction of the instantaneous phase of general CPM signals. Inaddition to the general advantages, as already mentioned above, ofdigital signal processing, the method has the advantage that the signalreconstruction can be carried out exactly for a choice of the functionsystem that is matched to the signal characteristics of theinstantaneous phase. This is only approximately the case with the normalmethods, for signal-theory reasons. Furthermore, a digital receiverbased on the method according to the invention allows an improvement inthe power efficiency, that is to say an improvement in the sensitivityand range for a predetermined maximum bit error rate.

[0020] In a further embodiment of the method according to the invention,the phase reconstruction may also be followed by a group delay timeequalizer for equalization of the group delay time equalization which iscaused by the analog channel selection filter.

[0021] Other features which are considered as characteristic for theinvention are set forth in the appended claims.

[0022] Although the invention is illustrated and described herein asembodied in a signal reception and processing method for cordlesscommunications systems, it is nevertheless not intended to be limited tothe details shown, since various modifications and structural changesmay be made therein without departing from the spirit of the inventionand within the scope and range of equivalents of the claims.

[0023] The construction and method of operation of the invention,however, together with additional objects and advantages thereof will bebest understood from the following description of specific embodimentswhen read in connection with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0024]FIG. 1 is a schematic circuit diagram of a receiver circuit whichoperates using the method according to the invention;

[0025]FIG. 2 is a schematic circuit diagram of a receiver circuit thathas been extended in comparison with FIG. 1; and

[0026]FIG. 3 is a graph plotting the scaling function for a Daubechieswavelet.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0027] Referring now to the figures of the drawing in detail and first,particularly, to FIG. 1 thereof, there is shown, by way of example, theconfiguration of a receiver circuit according to the invention which maybe used, for example, in DECT, WDCT, Bluetooth, SWAP, WLAN, IEEE802.11systems (frequency hopping method).

[0028] A radio signal is received by an antenna A and is supplied via aninput filter F1 to a low-noise input amplifier LNA. The input amplifierLNA amplifies the radio-frequency antenna signal with a variable gain.

[0029] After the low-noise amplification, the amplified signal isconverted to an intermediate frequency. For this purpose, the outputsignal from the low-noise amplifier LNA is supplied to two mixers M1 andM2. The mixers M1 and M2 are operated in a known manner with a phaseoffset of 90° using a mixing frequency which is derived from anon-illustrated local oscillator. The two signals which are used foroperation of the mixers M1 and M2 have a corresponding time relationshipcos(ω₀t) and sin(ω₀t), respectively. The term ω₀ refers to the angularfrequency associated with the oscillator frequency, and t is the time.

[0030] In-phase (I) and quadrature (Q) signals are produced at theoutputs of the mixers M1 and M2, respectively, at a reduced frequency,referred to in the following text as the intermediate frequency (IF).

[0031] The outputs from the two mixers M1 and M2 are suppliedrespectively to an I and a Q signal input of an analog channel selectionfilter KSF, which is used for mirror frequency suppression. The channelselection filter KSF is used to select a specific frequency channel, andhence to select the desired useful signal from the broadbandsignal/interference signal mixture which is present on the input side.

[0032] The two I and Q signal components are emitted, with the bandwidthof the useful channel, at two outputs A1, A2 of the channel selectionfilter KSF.

[0033] The output A1 of the channel selection filter KSF is connected toone input of a first limiter L1, and the output A2 is connected to oneinput of a second, physically identical, limiter L2.

[0034] The outputs of the limiters L1 and L2 are connected to respectiveinputs of a first and of a second sampling stage AS1 and AS2,respectively. The digital signal processing starts in the signal pathdownstream from the sampling stages AS1 and AS2.

[0035] The combination of limiters (L1 and L2, respectively) andsampling stages (AS1 and AS2, respectively) represents an analog/digitalconverter with a word length of 1. The method of operation of thiscombination of limiters and sampling stages, that is to say L1, AS1 andL2, AS2, is as follows:

[0036] The limiters, L1, L2 cut off all input levels above apredetermined limiter level threshold. In other words, in the cut-offrange, they produce an output signal with a constant signal level. If,as in the present case, the limiters L1, L2 have high gain and/or a lowlimiter level threshold, they are operating virtually all the time inthe cut-off or limiter range. A signal which has a discrete value(binary) that is still continuous in time is thus produced at the outputof the limiters L1, L2. The useful information in the I and Q signalcomponents at the outputs of the limiters L1 and L2 comprises the zerocrossings of these signal components.

[0037] The discrete-value analog signal components are sampled at a ratef_(s) by way of the two sampling stages AS1, AS2, which are in the formof one-bit samplers. The sampling is carried out with oversampling withrespect to the channel bandwidth (that is to say the bandwidth of thesignal downstream from the channel selection filter KSF).

[0038] By way of example, the channel bandwidth may be 1 MHz and thesampling frequency f_(s)=104 MHz. That is to say, oversampling by afactor of 104 can be carried out.

[0039] One advantage of this analog/digital conversion is that thelimiters L1, L2 suppress amplitude interference in the useful signal.

[0040] The digitized I and Q signal components are supplied to a phasereconstruction circuit PRS, in which the instantaneous phase φ(t) isreconstructed numerically using the zero crossings {t_(i)} and the phasevalues {φ(t_(i))=k_(i)·π/2, k_(i)εN₀} which can be determined with asuitably selected intermediate frequency are reconstructed using thefollowing reconstruction algorithm. In this case, s(t) is the signalwhich is to be reconstructed using an orthogonal function system{φ(t−k)}. $\begin{matrix}{{{Initialization}\quad {for}\quad k} = {{0\quad {to}\quad K} - 1}} \\{{c_{0,k} = {\sum\limits_{i = 0}^{I - 1}\quad {{s( t_{i} )} \cdot a_{i,k}}}}\quad} \\{{{for}\quad i} = {{0\quad {to}\quad I} - 1}} \\{a_{i,k} = {\int_{\frac{t_{i - 1} + t_{i}}{2} - k}^{\frac{t_{i} + t_{i + 1}}{2} - k}{{\phi (t)}\quad {t}}}} \\{{{Iteration}\quad {for}\quad n} = {{0\quad {to}\quad N} - 1}} \\{{{for}\quad k} = {{0\quad {to}\quad K} - 1}} \\{\quad {c_{{n + 1},k} = {c_{n,k} + {\sum\limits_{i = 0}^{I - 1}\quad {\lbrack {{s( t_{i} )} - {s_{n - 1}( t_{i} )}} \rbrack \cdot a_{i,k}}}}}} \\{{{for}\quad i} = {{0\quad {to}\quad I} - 1}} \\{{s_{n}( t_{i} )} = {\sum\limits_{k = 0}^{K - 1}{c_{n,k} \cdot {\phi ( {t_{i} - k} )}}}} \\{{{{Reconstruction}\quad {\hat{s}(t)}} = {\sum\limits_{k = 0}^{K - 1}{{c_{N,k} \cdot \phi}( {t - k} )}}}\quad}\end{matrix}$

[0041] By way of example, shifted orthogonal sinc functions ororthogonal scaling functions such as wavelets may be used for thisfunction system {φ(t−k)}. By way of example, FIG. 3 shows a Daubechieswavelength of length 6 for a scaling function. Daubechies scalingfunctions have the advantage of a finite carrier.

[0042] In order to improve the signal quality and for noise filtering,it is also possible, as shown in FIG. 1, to carry out postfiltering bymeans of a digital filter F2 with the system function H_(post)(z).

[0043]FIG. 2 shows an embodiment of a receiver circuit which is extendedin comparison to the embodiment shown in FIG. 1. In this receivercircuit, a group delay time equalizer is arranged downstream from thephase reconstruction circuit PRS, for equalization of the group delaydistortion that is caused by the analog channel selection filter. Thegroup delay time equalizer comprises all-pass filters AP1 and AP2, whichare arranged in the appropriate signal paths. The I and Q signaloutputs, respectively, of the all-pass filters AP1, AP2 may be suppliedto appropriate inputs of a suitable demodulator.

[0044] In the general case, the demodulator may be a CPM ContinuousPhase Modulation) demodulator. This uses the signal components which aresupplied to its inputs, that is to say the instantaneous phase or theinstantaneous frequency of these signal components, to estimate the datasymbols in the transmitted data symbol sequence.

We claim:
 1. A method for processing a signal, which comprises thefollowing steps: receiving the signal; carrying out channel selectionwith an analog channel selection filter; converting the signal to adigital, discrete-time and discrete-value signal; mathematicallyreconstructing a continuous-time and continuous-value signal profileusing zero crossings {t_(i)} and phase values {φ(t_(i))=k_(i)·π/2,k_(i)εN₀} by way of a mathematical reconstruction algorithm using afunction system {φ(t−k)}.
 2. The method according to claim 1, whereinthe receiving step comprises receiving a digitally modulated signal in acordless communications system.
 3. The method according to claim 1,wherein the function system is an orthogonal function system.
 4. Themethod according to claim 1, which comprises limiting the signal andoversampling the limited signal for digitizing the received signal. 5.The method according to claim 4, wherein the oversampling step comprisesproducing a signal with a word length of
 1. 6. The method according toclaim 1, which comprises FSK-modulating the signal.
 7. The methodaccording to claim 1, which comprises carrying out group delay timeequalization in a signal path downstream from the mathematicalreconstruction.
 8. The method according to claim 1, which comprisesconverting a signal frequency to an intermediate frequency after thechannel selection.
 9. A receiver circuit for a cordless communicationssystem, comprising: an analog signal processing section and a digitalsignal processing section; said analog signal processing sectioncontaining a channel selection filter; said digital signal processingsection containing a phase reconstruction circuit for mathematicalreconstruction of a continuous-time and continuous-value signal profileusing zero crossings {t_(i)} and periodic phase values{φ(t_(i))=k_(i)·π/2, k_(i)εN₀}, by way of a mathematical reconstructionalgorithm using a function system {φ(t−k)}.
 10. The receiver circuitaccording to claim 9, wherein said digital signal processing sectionincludes a group delay time equalizer for equalization of at least thesignal distortion caused by said channel selection filter.
 11. Thereceiver circuit according to claim 10, wherein said group delay timeequalizer is an all-pass filter.
 12. A receiver circuit for a cordlesscommunications system, comprising: an analog signal processing sectionand a digital signal processing section connected to said analog signalprocessing section; said analog signal processing section containing achannel selection filter; said digital signal processing sectioncontaining a phase reconstruction circuit programmed to process amathematical reconstruction algorithm using a function system {φ(t−k)}for mathematical reconstruction of a continuous-time andcontinuous-value signal profile using zero crossings {t_(i)} andperiodic phase values {φ(t_(i))=k_(i)·π/2, k_(i)εN₀}.
 13. The receivercircuit according to claim 12, wherein said digital signal processingsection includes a group delay time equalizer for equalization of atleast the signal distortion caused by said channel selection filter. 14.The receiver circuit according to claim 13, wherein said group delaytime equalizer is an all-pass filter.